Universal Power Conversion Methods

ABSTRACT

Methods and systems for transforming electric power between two or more portals. Any or all portals can be DC, single phase AC, or multi-phase AC. Conversion is accomplished by a plurality of bi-directional conducting and blocking semiconductor switches which alternately connect an inductor and parallel capacitor between said portals, such that energy is transferred into the inductor from one or more input portals and/or phases, then the energy is transferred out of the inductor to one or more output portals and/or phases, with said parallel capacitor facilitating “soft” turn-off, and with any excess inductor energy being returned back to the input. Soft turn-on and reverse recovery is also facilitated. Said bi-directional switches allow for two power transfers per inductor/capacitor cycle, thereby maximizing inductor/capacitor utilization as well as providing for optimum converter operation with high input/output voltage ratios. Control means coordinate the switches to accomplish the desired power transfers.

CROSS-REFERENCE TO OTHER APPLICATON

Priority is claimed from U.S. provisional application 60/811,191 filedJun. 6, 2006, which is hereby incorporated by reference.

BACKGROUND AND SUMMARY OF THE INVENTIONS

The present application relates to electric power conversion, and moreparticularly to buck-boost converter circuits, methods and systems whichcan convert DC to DC, DC to AC, and AC-AC, and are suitable forapplications including line power conditioners, battery chargers, hybridvehicle power systems, solar power systems, motor drives, and utilitypower conversion.

Numerous techniques have been proposed for electronic conversion ofelectric power from one form into another. A technique in commoncommercial usage for operating three phase induction motors at variablefrequency and voltage off of fixed frequency and voltage utility poweris the AC-DC-AC technique of the input diode bridge, DC-link capacitor,and the output active switch bridge, under PWM control, is shown in FIG.3. This motor drive technique (“standard drive”) results in compact andlow-cost motor drives, since no magnetic components are required andonly six active switches are needed.

A number of difficulties exist with the standard drive, however. Theinput current, while nominally in phase with the input voltage, istypically drawn in pulses. These pulses cause increased electric lossesin the entire electrical distribution system. The pulses also causehigher losses in the DC link capacitor. These losses reduce theefficiency of the drive, and also lessen the useful life of the DC linkcapacitor (commonly an Aluminum Electrolytic type), which has a limitedlife in an case. If the impedance of the source power is too low, thepulses may become so large as to be unmanageable, in which case it isnecessary to add reactance in the input lines, which increases losses,size, cost, and weight of the drive. Also, the voltage available for theoutput section is reduced, which may lead to loss-producing harmonics orlower-than-design voltage on the output waveform when full power, fullspeed motor operation is called for.

Due to the fixed DC-link voltage, the output switches are typicallyoperated with Pulse Width Modulation (PWM) to synthesize aquasi-sinusoidal current waveform into the motor, using the inductanceof the motor to translate the high voltage switched waveform from thedrive into a more sinusoidal shape for the current. While this doeseliminate lower order harmonics, the resulting high frequency harmonicscause additional losses in the motor due to eddy current losses,additional IR (ohmic) heating, and dielectric losses. These lossessignificantly increase the nominal losses of the motor, which reducesenergy efficiency, resulting in higher motor temperatures, which reducesthe useful life of the motor, and/or reduces the power available fromthe motor. Additionally, due to transmission line effects, the motor maybe subject to voltages double the nominal peak-to-peak line voltage,which reduces the life of the motor by degrading its insulation. Theapplied motor voltages are also not balanced relative to ground, and mayhave sudden deviations from such balance, which can result in currentflow through the motor bearings for grounded motor frames, causingbearing damage and reduced motor life. The sudden voltage swings at themotor input also cause objectionable sound emissions from the motor.

The output switches used in this motor drive must be constructed forvery fast operation and very high dV/dt in order to minimize lossesduring PWM switching. This requirement leads to selection of switcheswith drastically reduced carrier lifetimes and limited internal gain.This in turn decreases the conductance of each device, such that moresilicon area is required for a given amount of current. Additionally,the switches must be constructed to provide current limiting in theevent of output line faults, which imposes additional design compromiseson the switches which further increase their cost and losses.

Another problem with the standard drive is that the DC link voltage mustalways be less than the average of the highest line-to-line inputvoltages, such that during periods of reduced input voltage (such aswhen other motors are started across-the-line), the DC link voltage isinsufficient to drive the motor.

Yet another difficulty with the standard drive is its susceptibility toinput voltage transients. Each of the input switches must be able towithstand the full, instantaneous, line-to-line input voltage, or atleast the voltage after any input filters. Severe input transients, asmay be caused by lightning strikes, may produce line-to-line voltagesthat exceed 2.3 times the normal peak line-to-line voltages, even withsuitable input protection devices such as Metal Oxide Varistors. Thisrequires that the switches be rated for accordingly high voltages (e.g.1600 volts for a 460 VAC drive), which increases cost per ampere ofdrive.

The standard drive also cannot return power from the DC link to theinput (regeneration), and therefore large braking resistors are requiredfor an application m which the motor must be quickly stopped with alarge inertial or gravitational load.

Modifications to the basic motor drive described above are available, asalso shown in FIG. 3, but invariably result in much higher costs, size,weight and losses. For example, in order to reduce input currentharmonics (distortion) and to allow for regeneration, the diode bridgemay be replaced by an active switch bridge identical to the outputswitch bridge, which is accompanied by an input filter consisting ofinductors and capacitors, all of which result in higher costs and drivelosses. Also, as shown in FIG. 3, output filters (“sine filter”) areavailable to change the output voltage waveform to a sinusoid, but againat the expense of greater cost, size, weight, and losses.

AC-AC line conditioners are constructed in a similar fashion to thestandard drive with input and output filters and an active front end,and also suffer from the above mentioned problems.

Other motor AC-AC converters are known, such as the Matrix Converter,Current Source Converter, or various resonant AC and DC link converters,but these either require fast switching devices and substantial inputand/or output filters, or large, lossy, and expensive reactivecomponents, or, as in the case of the Matrix Converter, are incapable ofproviding an output voltage equal to the input voltage.

The term “converter” is sometimes used to refer specifically to DC-to-DCconverters, as distinct from DC-AC “inverters” and AC-AC“cycloconverters.” However, in the present application the wordconverter is used more generally, to refer to all of these types andmore.

What is needed then is a converter technique which draws power from theutility lines with low harmonics and unit power factor, is capable ofoperating with full output voltage even with reduced input voltage,allows operations of its switches with low stress during turn-off andturn-on, is inherently immune to line faults, produces voltage andcurrent output waveforms with low harmonics and no common mode offsetswhile accommodating all power factors over the full output frequencyrange, operates with high efficiency, and which does so at a reasonablecost in a compact, light-weight package.

DC-DC, DC-AC, and AC-AC Buck-Boost converters are shown in the patentand academic literature which have at least some of the aforementioneddesirable attributes. The classic Buck-Boost converter operates theinductor with continuous current, and the inductor may have an input andoutput winding to form a transformer for isolation and/orvoltage/current translation, in which case it is referred to as aFlyback Converter. There are many examples of this basic converter, allof which are necessarily hard switched and therefore do not have thesoft-switched attribute, which leads to reduced converter efficiency andhigher costs. An example of a hard switched 3 phase to 3 phaseBuck-Boost converter is shown in FIG. 4, from K. Ngo, “Topology andAnalysis in PWM Inversion, Rectification, and Cycloconversion,”Dissertation, California Institute of Technology (1984).

One proposed DC-AC Buck-Boost converter (in U.S. Pat. No. 5,903,448)incorporates a bi-directional conduction/blocking switch in its outputsection to accommodate four quadrant operation, with AC output andbi-directional power transfer. The input, however, cannot be AC, and ituses hard switching.

Universal Power Converter

The present application discloses new approaches to power conversion. Alink reactance is connected to switching bridges on both input andoutput sides, and driven into a full AC waveform.

In some preferred embodiments (but not necessarily in the link reactanceis driven with a nonsinusoidal waveform, unlike resonant converters.

In some preferred embodiments (but not necessarily in all), capacitivereactances are used on both input and output sides.

In some preferred embodiments (but not necessarily in all), theswitching bridges are constructed with bidirectional semiconductordevices, and operated in a soft-switched mode.

In some preferred embodiments (but not necessarily in all), the inputswitching bridge is operated to provide two drive phases, from differentlegs of a polyphase input, during each cycle of the link reactance. Theoutput bridge is preferably operated analogously, to provide two outputconnection phases during each cycle of the reactance.

In some preferred embodiments (but not necessarily in all), the linkreactance uses an inductor which is paralleled with a discretecapacitor, or which itself has a high parasitic capacitance.

The disclosed innovations, in various embodiments, provide one or moreof at least the following advantages:

-   -   a high-bandwidth active control ability—more so than resonant or        voltage-source or current-source converters    -   Design versatility    -   Power efficiency    -   Optimal use of device voltage ratings    -   High power density converters    -   High power quality (low input and output harmonics with minimal        filtering)    -   Voltage buck and boost capability    -   Bi-directional, or multi-directional power transfer capability    -   High frequency power transformer capability, allowing for        compact active transformer and full galvanic isolation if        desired.    -   Input-Output isolation even without a transformer, allowing for        output with no common-mode voltage    -   Moderate parts count resulting from absence of auxiliary power        circuits for snubbing    -   High-bandwidth active control ability—more so than resonant or        voltage-source or current-source converters

BRIEF DESCRIPTION OF THE DRAWINGS

The disclosed inventions will be described with reference to theaccompanying drawings, which show important sample embodiments of theinvention and which are incorporated in the specification hereof byreference. These drawings illustrate by way of example and notlimitation.

FIG. 1 shows a sample embodiment as a Full-Bridge Buck-Boost Converterin a Three Phase AC Full Cycle Topology with Bi-directional Conductingand Blocking Switches (BCBS). Each BCBS is shown as appears in FIG. 2 ofU.S. Pat. No. 5,977,569, also shown as switch 201 of FIG. 2. Inputfilter capacitors 130 are placed between the input phases and outputfilter capacitors 131 similarly attached between the output phases inorder to closely approximate voltage sources and to smooth the currentpulses produced by the switches and the inductor 120. Output filtercapacitors are preferably attached in a grounded Y configuration asshown. An input line reactor 132 may be needed in some applications toisolate the voltage ripple on the input capacitors 130 from the utility122.

FIGS. 2 a-2 d show four alternative versions of the basic Bi-directionalConducting and Blocking Switch (BCBS). FIG. 2 a is an anti-parallel pairof commercially available Reverse-Blocking IGBTs (IXRH 40N120, 1200Volt, 55 A). FIG. 2 b is the switch cited in U.S. Pat. No. 5,977,569.FIG. 2 c is an anti-parallel pair of commercially available IGBTs inseries with diodes. FIG. 2 d is an anti-parallel pair of commerciallyavailable GTOs. Many other BCBS switch configurations are possible. EachBCBS can block voltage and conduct current in either direction.

FIG. 3 shows Prior Art for the “Standard Drive”, which is the mostcommon low voltage motor drive type available, and is a voltage sourcepulse width modulated (PWM) topology. Also shown are various options toallow this drive to achieve more acceptable operation.

FIG. 4 shows a conventional hard-switched three phase to three phase ACbuck-boost converter.

FIG. 5 shows a conventional soft-switched “partial resonant” three phaseto three phase AC buck-boost converter, which has uni-directionalswitches, suffers from 1) a long quiescent resonant “swing back” timewhen no power is transferred, 2) greatly reduced frequency of operationas the voltage ratio between input and output is increased, and 3)inability to sink or source output current as the output voltageapproaches zero.

FIG. 6 shows the inductor current and voltage waveforms for theconverter, including the “swing back” of inductor voltage in time periodM4.

FIG. 7 shows a prior art resonant link inverter that bears a superficialresemblance to this invention, but which in fact is completelydifferent, as the voltage is sinusoidal and double the nominal inputvoltage, and the switches serve only to connect positive or negativelink half-cycles to the input and output.

FIG. 8 show Prior Art of U.S. Pat. No. 7,057,905 which has somesimilarities to this invention in that it is a buck-boost converterwhich uses bi-directional switches. It is, however, basically aconventional, hard-switched, buck-boost converter which usesbi-directional switches to allow it operate with a DC component in theinductor in either direction.

FIG. 9 shows the input line voltages for the current switching exampleof FIGS. 11, 12 and 13, with the phase designations corresponding tothose of FIG. 1.

FIG. 10 shows the output line voltages for the current switching exampleof FIGS. 11, 12 and 13, with the phase designations corresponding tothose of FIG. 1.

FIG. 11 summarizes the line and inductor current waveforms for a fewinductor cycles at and around the inductor cycle of FIGS. 12 and 13.

FIGS. 12 a-12 j show voltage and current waveforms on the inductorduring a typical cycle while transferring power at full load from inputto output, as occurs in FIG. 5 while operating the motor at full power,including full output voltage. FIGS. 12 b and 12 g are used to summarizeinductor/capacitor voltage ramping current flow between modes, with 12 bshowing it for positive inductor current, and 12 g for negative inductorcurrent. When minimum voltage phase pairs are mentioned, that refers tophase pairs with opposing current, not necessarily the minimum voltagephase pair, as that is often a phase pair with current going in the samedirection.

FIG. 13 shows voltage and current waveforms corresponding to the fullpower condition of FIG. 12, with the conduction mode numberscorresponding to the mode numbers of FIG. 12.

FIG. 14 is similar to FIG. 13, but shows inductor voltage and currentfor an output voltage of about half the full output voltage.

FIG. 15 shows an embodiment of the invention with the Full Bridge ThreePhase Cycle Topology, with Controls and I/O Filtering, including a threephase input line reactor as needed to isolate the small but highfrequency voltage ripple on the input filter capacitors from theutility.

FIG. 16 illustrates current and timing relationships for a DC or singlephase AC converter with output voltage equal to the input voltage.

FIG. 17 shows the same current and timing relationships as FIG. 16, butwith the output voltage ½ of the input voltage.

FIG. 18 is a spreadsheet, with equations as shown, that calculates theaverage output current for a given set of conditions, as the currentdischarge time is varied. These equations may be used in a controlsystem to control switch timing to give a commanded output current.

FIG. 19 shows the results of the spreadsheet of FIG. 18 for the statedconditions with four output voltages as a function of output dischargetime. Also noted on the curves are the inductor operating frequency.

FIG. 20 is a version of FIGS. 16 and 17 which shows inductor current andtiming for a regeneration condition where the output voltage is ½ of theinput.

FIG. 21 shows an embodiment of the invention with the DC or Single Phaseportals. If one of the portals is DC, and is always a higher voltagethan the other portal, one-way blocking switches may be used on thatportal.

FIG. 22 shows an embodiment of the invention with aTransformer/Inductor, as is common with other Buck-Boost converters inthe Flyback configuration. Any of the embodiments of this invention mayuse a transformer/inductor in place of the inductor if full isolationand/or voltage and current transforming is desired. Even without thetransformer, just using the inductor, a degree of isolation is providedsince the input and output lines are never directly connected together.If a transformer/inductor is used, the transformer must have at leastsome air gap, in order to produce a magnetizing inductance that does notsaturate at the peak current that is used.

FIG. 23 shows an embodiment of the invention in a four portalapplication mixing single phase AC and multiple DC portals, as may beused to advantage in a Solar Power application. Other topologies mustuse at least two separate converters to handle this many portals. Theswitches attached to the solar power source need only be one wayswitches, since power the source is DC and power can only be transferredout of the device. The switches could even be non-reverse blocking ifthe DC output only source was always guaranteed to be higher voltagethan all other voltage sources.

FIG. 24 shows an embodiment of the invention in a three portalapplication mixing three phase AC portals and a DC portal, as may beused to advantage in a Hybrid Electric Vehicle application.

FIG. 25 shows an embodiment as a Half-Bridge Buck-Boost Converter in aSingle Phase AC or DC Topology with BCBS. The half bridge topologyrequires half as many switches, but also results in half the powertransfer for equivalent switch ratings, and higher per unit ripplecurrent in the input and output filters

FIG. 26 show a sample embodiment in a Half-Bridge Buck-Boost Converterin a Three Phase AC Topology with BCBS. Again, the half bridge topologyrequires half as many switches, but also results in half the powertransfer for equivalent switch ratings, and higher per unit ripplecurrent in the input and output filters

FIG. 27 shows a sample embodiment in a single phase to three phasesynchronous motor drive.

FIG. 28 shows a sample embodiment with dual, parallel, “power modules”,each of which consists of 12 bi-directional switches and a parallelinductor/capacitor. More than two power modules may of course be usedfor additional options in multiway conversion.

FIG. 29 shows an embodiment of this invention as a three phase PowerLine Conditioner, in which role it may act as an Active Filter and/orsupply or absorb reactive power to control the power factor on theutility lines.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

The numerous innovative teachings of the present application will bedescribed with particular reference to presently preferred embodiments(by way of example, and not of limitation).

Contrast with Other Approaches

DC-DC Buck-Boost converters employing resonant techniques to achievesoft switching are also shown in the patent literature (examples areU.S. Pat. Nos. 4,616,300, issued Oct. 7, 1986; 6,404,654, issued Jun.11, 2002). These are not capable of DC-AC or AC-AC operation, and arealso limited in their DC-DC range, in that the output DC voltage must belarger than some minimum in order to achieve zero voltage turn-on of thepower switch. In contrast to this prior art, the inventions describedbelow have no restrictions on the relative voltages between the inputand output portals, and power transfer is bi-directional.

A “partial-resonant” 3 phase AC-AC Buck-Boost converter is described inKim et al., “New Bilateral Zero Voltage Switching AC/AC Converter UsingHigh Frequency Partial-resonant Link”, Korea Advanced Institute ofScience and Technology, (IEEE 1990), and shown in FIG. 5, which usesuni-directional switches. This converter has many desirable attributes,including soft-switching, but has important differences from theinventive circuits and methods described below:

-   -   1) has significantly reduced utilization of the        inductor/capacitor,    -   2) has higher per unit RMS current loading on the input/output        capacitors,    -   3) has a lower operating frequency for a given turn-off        condition which leads to larger, costlier, and less efficient        I/O filtering,    -   4) cannot deliver or receive current to/from the output for        sufficiently low output voltages and/or power factors,    -   5) and has no lower limit on the operating frequency as output        power factor and/or output voltage approaches zero.        The lowered operating frequency can lead to destructive        resonances with the required input filters. Input filters were        not shown in this reference, but are normally required. As shown        in FIG. 6 (also from the Kim et al. paper), time period M4,        reduced inductor/capacitor utilization results from the resonant        “swing back” time as the voltage on the inductor/capacitor        resonantly swings from the output voltage back to the input        voltage, and at full power typically requires 33% of the total        power cycle, such that no power transfer occurs for 33% of the        time. Thus, that converter achieves only one power transfer        cycle for each cycle of the inductor, whereas the converter of        FIG. 1 preferably has two power transfers per inductor cycle, as        enabled by the use of bi-directional switches.

FIG. 7 shows another prior art converter, from Rajashekara et al.,“Power Electronics”, Chapter 30 in The Electrical Engineering Handbook(ed. R. Dorf 2000). That converter bears a superficial resemblance tothe converter of FIG. 1, in that it has 12 bi-directional switches and aparallel inductor/capacitor; but the topology is different, and the modeof operation is totally different. The converter of FIG. 7 does not haveI/O filter capacitors, and indeed cannot operate with such capacitors.The converter of FIG. 7 is actually a resonant link converter, such thatthe inductor/capacitor voltage and current is sinusoidal and resonant asshown in the figure. That converter must be isolated from voltagesources and sinks by inductance (e.g. line reactors, transformers, ormotor inductance), since the voltages between its switches and saidinductance rapidly swing over a range almost twice as high as the peakline-to-line voltages. (Various such high voltages are imposed on theinput and output inductances by selectively enabling/disablingappropriate switch pairs for each half cycle of the inductor/capacitor,as indicated.) By contrast, the converter of FIG. 1 is not resonant, andthe peak inductor voltage is just the peak line-to-line voltage. Theconverter of FIG. 1 could not be operated as the converter of FIG. 7,and the converter of FIG. 7 could not be operated as the invention ofFIG. 1.

U.S. Pat. No. 7,057,905 shows a buck-boost power converter withbi-directional switches and a method of operating same. This is aconventional hard-switched buck-boost converter, in that it has nocapacitance in parallel with the inductor and has only one power cycleper inductor cycle, except that the additional input switch capabilityallows it to operate with an inductor DC offset current in eitherdirection. It may also apply both polarities to the inductor during asingle power cycle to better control the operating frequency.

As compared with this invention, U.S. Pat. No. 7,057,905, operates witha DC bias current in the inductor, such that it cannot do two powercycles per inductor cycle as this invention can, and cannot therefore dosoft switching. It is prohibited from doing so since, as shown in FIG.8, it only has the two output switches 80 and 81, such that, in order totransfer current to the output in one direction, the inductor currentmust also be in that same direction. Thus, in order to produce currentflow into the output capacitor 24, current must flow “upwards” throughinductor 82. To replenish inductor energy transferred to the output, theinductor must be reconnected to the input with opposite polarity, whichis a hard switched operation, necessitating the hard reverse turn-off ofoutput switches 80 and 81 by turning on two appropriate input switches.This invention, in contrast, simply turns off the two output switches,which then causes the inductor voltage to increase to the input level,but with opposite polarity from the previous input connection, andexcess inductor energy (if any) is returned back to the input, and theninductor current is reversed, after which inductor energy is againtransferred to the output, but with opposite current flow, facilitatedby additional bi-directional switches that connect the inductor to theoutput with opposite polarity, said switches not being present in U.S.Pat. No. 7,057,905. Additionally, since in this invention the inductorvoltage is never forced from its “natural” direction, capacitance inparallel with the inductor is allowed to facilitate soft turn-off. Saidsoft switching of this invention allows this invention to operate withmuch higher switching frequencies with consequent large reductions inthe reactive component sizes and losses.

The prior art of Buck-Boost resonant converters are not capable ofoperating, as this invention does, in the “Full Cycle” mode (describedbelow), in which the inductor (or transformer) is operated with fullalternating current, with no DC component in any windings. This mode ofoperation requires bi-directional (AC) switches, and produces two powertransfers for each cycle of the inductor/capacitor, resulting insuperior utilization of the inductor/capacitor and I/O filters, whilealso allowing current transfer at low output voltages or low powerfactors.

The shortcomings described above are not intended to be exhaustive, butrather among the many that tend to impair the effectiveness ofpreviously known techniques for power conversion. Other noteworthyproblems may also exist; however, those mentioned here are sufficient todemonstrate that methodologies appearing in the art have not beenaltogether satisfactory.

Highlights and Overview

The shortcomings listed above are reduced or eliminated by the disclosedtechniques. These techniques are applicable to a vast number ofapplications, including but not limited to all DC-DC, DC-AC, and AC-ACpower conversions.

The present application discloses power converters which are generallyof the Buck-Boost family, but which use capacitance, either parasiticalone or with added discrete device(s), in parallel with the Buck-Boostinductor to achieve low turn-off switching stresses (i.e. “softswitching”) on the semiconductor switches, allowing relatively slow andinexpensive switches to be used. In alternative disclosed embodiments,as discussed below, operation without such added capacitance ispossible, at the expense of higher forward turn-off switching losses.The converter of FIG. 5 cannot operate without the parallel capacitor,as it would then become the classic hard-switched buck-boost converterof Ngo.

In FIG. 1, and various other disclosed embodiments, even with little orno parallel capacitance is used, switch turn on always occurs as theswitch transitions from reverse to forward bias, allowing for lowturn-on losses. Reverse recovery of the switches is accomplished withlow rates of current decrease, and with low reverse recovery voltage,leading to near zero loss reverse recovery switching.

The embodiments described below are believed to be the first applicationof the Buck-Boost inductor in flail Alternating Current (AC) mode, whichis referred to herein as the “Full Cycle” mode and which results in twopower transfers per inductor cycle. Buck-Boost converters, includingthose of the Ngo and Kim references cited above, have a DC bias in theinductor current, and only one power transfer per inductor cycle.

The disclosed inventions can also be used for DC-AC, AC-DC, AC-AC, orDC-DC conversion, with no limitation on the relative magnitudes of thevoltages involved as long as the voltage rating of the switches is notexceeded. However, if the implementation is such that one portal isalways a higher voltage than the other portal, then the switchesconnected to said higher portal need only be able to block voltage inone direction.

Full electrical isolation and/or greater voltage and current conversionmay be achieved by using an inductor/transformer instead of the simpleinductor. Note that the inductor/transformer will typically not havecurrent in both sides at the same time, so its operation is more like asplit inductor (as in a flyback converter) than like a simpletransformer (as in a push-pull converter. Another significant differencebetween buck-boost and push-pull is that the push-pull output voltage isfixed as a multiple or fraction of the input voltage, as given by theturns ratio, while the buck-boost has no such limitation. Push-pulltopologies are described athttp://en.wikipedia.org/wiki/Push-Pull_Converter, which (in its state asof the filing date) is hereby incorporated by reference. A push-pull isquite unlike a buck-boost or flyback converter, in that the transformeris not operated as an energy-transfer inductor. In a buck-boost orflyback, input current pumps energy into a magnetic field, which is thendrained to drive output current; thus the input and output currents flowat different times.

inductor/transformer leakage inductance is typically a significantconcern of buck-boost designs. This is typically dealt with byminimizing the leakage, and sometimes by adding circuit elements to dealwith it. By contrast, the inventions described below can tolerate largeparasitic capacitance, and thus inductors or transformers with veryclose windings can be specified, to minimize the leakage inductance. Thestandard hard switched buck-boost cannot tolerate parasitic capacitance,which makes it very difficult to minimize the leakage inductance forthose configurations.

The innovative converter circuits, in various embodiments areconstructed of semiconductor switches, an inductor, advantageously acapacitor in parallel with the inductor, and input and output filtercapacitances. A control means, controlling the input switches, firstconnects the inductor, initially at zero current, to the input voltage,which may be DC or the highest line-to-line voltage AC pair in a threephase input, except at startup, in which case a near zero-voltage linepair is used. The control then turns off those switches when the currentreaches a point determined by the control to result in the desired rateof power transfer. The current then circulates between the inductor andcapacitor, which results in a relatively low rate of voltage increase,such that the switches are substantially off before the voltage acrossthem has risen significantly, resulting in low turn-off losses.

With DC or single phase AC input, no further current is drawn from theinput. With 3 phase AC input, the control will again connect theinductor to the input lines, but this time to the line-to-line pairwhich has a lower voltage then the first pair. Turn on is accomplishedas the relevant switches transition from reverse to forward bias. Afterdrawing the appropriate amount of charge (which may be zero if thecontrol determines that no current is to be drawn from the pair, as forexample that the pair is at zero volts and input unity power factor isdesired), the relevant switches are again turned off. Under mostconditions, the voltage on the inductor will then reverse (withrelatively low rates of voltage change due to the parallel capacitance).With 3 phase AC output, the control will turn on switches to allowcurrent to flow from the inductor to the lowest voltage pair of lineswhich need current, after the relevant switches become forward biased,with the control turning off the switches after the appropriate amountof charge has been transferred. The inductor voltage then ramps up tothe highest output line-to-line pair for 3 phase AC, or to the outputvoltage for single phase AC or DC. Again, switches are turned on totransfer energy (charge) to the output, transitioning from reverse toforward bias as the voltage ramps up. If the output voltage is largerthen the highest input voltage, the current is allowed to drop to zero,which turns off the switch with a low rate of current reduction, whichallows for the use of relatively slow reverse recovery characteristics.If the output voltage is less then the highest input voltage, theswitches are turned off before current stops, so that the inductorvoltage ramps up to the input voltage, such that zero-voltage turn on ismaintained. Alternatively, the switches may be turned off before thepoint cited in the previous sentence, so as to limit the amount ofcurrent into the output. In this case, the excess energy due to currentin the inductor is directed back into the input by turning on switchesto direct current flow from the inductor into either e highest voltagepair in three phase, or the single phase AC or DC input.

In a three phase AC converter, the relative charge per cycle allocatedto each input and output line pair is controlled to match the relativecurrent levels on each line (phase). After the above scenario, when zerocurrent is reached the inductor is reconnected to the input, but with apolarity reversed from the first connection, using switches that arecomplimentary to the switches used in the first half of the cycle. Thisconnection can occur immediately after zero current (or shortly afterzero current if the input voltage is less than the output voltage, toallow the capacitor voltage time to ramp back down), giving fullutilization of the power transfer capability of the inductor. Noresonant reversal is required as in the time period M4 of the Kimconverter shown in FIGS. 5 and 6.

The disclosed embodiments are inherently capable of regeneration at anycondition of output voltage, power factor, or frequency so in motordrive or wind power applications, the motor may act as a generator,returning power to the utility lines.

In an AC motor drive implementation, input and output filtering may beas little as line-to-neutral connected capacitors. Since switchinglosses are very low due to soft switching, the Buck-Boost inductor canbe operated at a high inductor frequency (typically 5 to 20 kHz for lowvoltage drives), allowing for a single, relatively small, and low loss,magnetic device. The current pulse frequency is twice the inductorfrequency. This high frequency also allows the input and output filtercapacitors to be relatively small with low, high frequency ripplevoltage, which in turns allows for small, low loss line reactors.

Input voltage “sags”, as are common when other motors are connectedacross the line, are accommodated by temporarily drawing more currentfrom the input to maintain a constant power draw and output voltage,utilizing the boost capability of this invention, avoiding expensiveshutdowns or even loss of toque to the application.

The full filter between the converter and an attached voltage source(utility) or sink (motor, another utility, or load) includes the linecapacitance (line-to-line or line-to-neutral, as in Y or Delta), and aseries line inductance (or hue reactor as It generally called). Whendriving a motor, the line reactance is just the inductance of the motor.I show this L-C filter in my preferred embodiments, and also mentionedit in my earlier claims. So it is a power filter, AND it does importantconditioning for the converter.

The preferred converter benefits from having very low impedence voltagesources and sinks at the inputs and outputs. (This is a significantdifference from the converter of FIG. 7, which has line reactance(inductors) at the I/O, not capacitance.) The link inductor current mustbe able to be very rapidly switched between the link capacitor and theI/O capacitors, and line reactance would prevent that from incurring,and in fact would likely destroy the switches. The physical constructionof the converter should preferably be carefully done to minimize allsuch inductance which may impair link reactance switching.

The line capacitance itself does not have to be really any particularvalue, but for proper operation the change in voltage on the linecapacitance while charging or discharging the link inductance shouldonly be a small fraction of the initial voltage, let's say less than10%. There are other restraints as well. For a 20 hp, 460 VAC prototype,80 microF of line-to-neutral capacitance results in only a 1 to 2%ripple voltage. (This large capacitance was chosen in order to get theripple current within the capacitor's current rating.) Capacitors couldbe made with lower uF for the same current rating, resulting in smaller,cheaper capacitors, and higher voltage ripple, but this is all that isavailable right now.

Another important consider is the resonant frequency formed by the LC ofthe line reactance and the line capacitance (the I/O power filter). Thisfrequency must be lower than the link power cycle frequency in order tonot have that filter resonant with the voltage ripple on the linecapacitance. For my 20 hp 460 VAC prototype, the link frequency is 10kHz, so the link power cycle frequency is 20 kHz (2 power cycles perlink voltage cycle), and the resonant frequency of the L-C I/O is lowerthan 2 kHz, so that works well.

So, to summarize, the capacitance needs to be large enough to reasonablystabilize the I/O voltage to allow the link inductor charge/discharge tooccur properly, and the L-C resonant frequency needs to be less thantwice the link voltage frequency, and generally at least 4 to 10 timeslower.

It should also be noted that too much capacitance on line filter canlead to excess reactive power on the utility connection.

Further Detail

Referring initially to FIG. 1, illustrated is a schematic of a threephase converter 100 that embodies the present invention. The converter100 is connected to a first and second power portals 122 and 123 each ofwhich may source or sink power, and each with a port for each phase ofthe portal. It is the function of said converter 100 to transferelectric power between said portals while accommodating a wide range ofvoltages, current levels, power factors, and frequencies between theportals. Said first portal may be for example, a 460 VAC three phaseutility connection, while said second portal may be a three phaseinduction motor which is to be operated al variable frequency andvoltage so as to achieve variable speed operation of said motor. Thisinvention may also accommodate additional portals on the same inductor,as may be desired to accommodate power transfer to and from other powersources ardor sinks, as shown in FIGS. 23 and 24.

Referring to FIG. 1, the converter 100 is comprised of a first set ofelectronic switches S_(1u), S_(2u), S_(3u), S_(4u), S_(5u), and S_(6u)that are connected between a first port 113 of a link inductor 120 andeach phase, 124 through 129, of the input portal, and a second set ofelectronic switches S_(1l), S_(2l), S,_(3l), S_(4l), S_(5l), and S_(6l)that are similarly connected between a second port 114 of link inductor120 and each phase of the output portal. A link capacitor 121 isconnected in parallel with the link inductor, forming the linkreactance. Each of these switches is capable of conducting current andblocking current in both directions, and may be composed of thebi-directional ICIBT 201 of FIG. 2, as shown in U.S. Pat. No. 5,977,569.Many other such bi-directional switch combinations are possible, such asanti-parallel reverse blocking IGBTs 200 of FIG. 2.

Most of these switch combinations contain two independently controlledgates, as shown with all the switches for FIG. 2, with each gatecontrolling current flow in one direction. In the following description,it is assumed that two gate switches are used in each switch, and thatthe only gate enabled in a switch is the gate which controls current inthe direction which is desired in the subsequent operation of theswitch. Thus, when each switch mentioned below is said to be enabled,said enabling occurs before conduction occurs, since that portion of theswitch is reversed biased at the instant of being enabled, and does notconduct until it becomes forward biased as a result of the changingvoltage on the parallel pair of inductor and capacitor. Any switchembodiment which has only one gate, such as a one-way switch embeddedwithin a full wave bridge rectifier, must be enabled only when thevoltage across it is very small, which requires precise and accuratetiming that may be difficult to achieve in practice.

The converter 100 also has input and output capacitor filters 130 and131, respectively, which smooth the current pulses produced by switchingcurrent into and out of inductor 120. Optionally, a line reactor 132 maybe added to the input to isolate the voltage ripple on input capacitorfilter 131 from the utility and other equipment that may be attached tothe utility lines. Similarly, another line reactor, not shown, may beused on the output if required by the application.

For illustration purposes, assume that power is to be transferred in afull cycle of the inductor/capacitor from the first to the secondportal, as is illustrated in FIG. 13. Also assume that, at the instantthe power cycle begins as shown in FIG. 9, phases A_(i) and B_(i) havethe highest line to line voltage of the first (input) portal, linkinductor 120 has no current, and link capacitor 121 is charged to thesame voltage as exists between phase A_(i) and B_(i). The controllerFPGA 1500, shown in FIG. 15, now turns on switches S_(1u) and S_(2l).whereupon current begins to flow from phases A_(i) and B_(i) into linkinductor 120, shown as Mode 1 of FIG. 12 a. FIG. 13 shows the inductorcurrent and voltage during the power cycle of FIG. 12, with theConduction Mode sequence 1300 corresponding to the Conduction Modes ofFIG. 12. The voltage on the link reactance remains almost constantduring each mode interval, varying only by the small amount the phasevoltage changes during that interval. After an appropriate current levelhas been reached, as determined by controller 1500 to achieve thedesired level of power transfer and current distribution among the inputphases, switch S_(2l) is turned off. Current now circulates, as shown inFIG. 12 b, between link inductor 120 and link capacitor 121, which isincluded in the circuit to slow the rate of voltage change, which inturn greatly reduces the energy dissipated in each switch as it turnsoff. In very high frequency embodiments of this invention, the capacitor121 may consist solely of the parasitic capacitance of the inductorand/or other circuit elements.

To continue with the cycle, as shown as Mode 2 FIG. 6 c and FIG. 13,switch S_(3l) is next enabled, along with the previously enabled switchS_(1u). As soon as the link reactance voltage drops to just less thanthe voltage across phases A_(i) and C_(i), which are assumed for thisexample to be at a lower line-to-line voltage than phases A_(i) andB_(i), as shown in FIG. 9, witches S_(1u) and S_(3l) become forwardbiased and start to further increase the current flow into the linkinductor, and the current into link capacitor temporarily stops. The two“on” switches, S_(1u) and S_(3l), are turned off when the desired peaklink inductor current is reached, said peak link inductor currentdetermining the maximum energy per cycle that may be transferred to theoutput. The link inductor and link, capacitor then again exchangecurrent, as shown if FIG. 12 b, with the result that the voltage on thelink reactance changes sign, as shown in graph 1301, between modes 2 and3 of FIG. 13. Now as shown in FIG. 12 d, output switches S_(5u) andS_(6l) are enabled, and start conducting inductor current into the motorphases A_(o) and B_(o), which are assumed in this example to have thelowest line-to-line voltages at the present instance on the motor, asshown in FIG. 10. After a portion of the inductor's energy has beentransferred to the load, as determined by the controller, switch S_(5u)is turned off, and S_(4u) is enabled, causing current to flow again intothe link capacitor, which increases the link inductor voltage until itbecomes slightly greater than the line-to-line voltage of phases A_(o)and C_(o), which are assumed in this example to have the highestline-to-line voltages on the motor, as shown in FIG. 10. As shown inFIG. 12 e, most of the remaining link inductor energy is thentransferred to this phase pair (into the motor), bringing the linkinductor current down to a low level. Switches S_(4u) and S_(6l) arethen turned off, causing the link inductor current again to be shuntedinto the link capacitor, raising the link reactance voltage to theslightly higher input line-to-line voltage on phases A_(i) and B_(i).Any excess link inductor energy is returned to the input. The linkinductor current then reverses, and the above described link reactancecurrent/voltage half-cycle repeats, but with switches that arecomplimentary to the first half-cycle, as is shown in FIGS. 6 f to 6 j,and in Conduction Mode sequence 1300, and graphs 1301 and 1302. FIG. 12g shows the link reactance current exchange during the inductor'snegative current half-cycle, between conduction modes.

FIG. 11 summarizes the line and inductor current waveforms for a fewlink reactance cycles at and around the cycle of FIGS. 12 and 13.

Note that TWO power cycles occur during each link reactance cycle: withreference to FIGS. 12 a-12 i, power is pumped IN during modes 1 and 2,extracted OUT during modes 3 and 4, IN again during modes 5 and 6, andOUT again during modes 7 and 8. The use of multi-leg drive produceseight modes rather than four, but even if polyphase input and/or outputis not used, the presence of TWO successive in and out cycles during onecycle of the inductor current is notable.

As shown in FIG. 12 and FIG. 13, Conduction Mode sequence 1300, and ingraphs 1301 and 1302, the link reactance continues to alternate betweenbeing connected to appropriate phase pairs and not connected at all,with current and power transfer occurring while connected, and voltageramping between phases while disconnected (as occurs between the closelyspaced dashed vertical lines of which 1303 in FIG. 13 is one example.

In general, when the controller 1500 deems it necessary, each switch isenabled, as is known in the art, by raising the voltage of the gate 204(FIG. 2) on switch 200 above the corresponding terminal 205, as anexample. Furthermore, each switch is enabled (in the preferred two gateversion of the switch) while the portion of the switch that is beingenabled is zero or reverse biased, such that the switch does not startconduction until the changing link reactance voltage causes the switchto become forward biased. Single gate AC switches may be used, as with aone-way switch embedded in a four diode bridge rectifier, but achievingzero-voltage turn on is difficult, and conduction losses are higher.

in FIG. 15, current through the inductor is sensed by sensor 1510, andthe FPGA 1500 integrates current flows to determine the current flowingin each phase (port) of the input and output portals. Phase voltagesensing circuits 1511 and 1512 allow the FPGA 1500 to control whichswitches to enable next, and when.

By contrast, note that the prior art structure of FIG. 8 has fourbi-directional switches on the input, and two on the output, with a linkinductor (no parallel capacitor) in between. That patent is a hardswitched buck-boost, and, like all prior buck-boost converters, it hasonly 1 power transfer per link inductor cycle. Moreover, the linkinductor has a DC current component, unlike the converter of FIG. 1(which has NO average DC current, only AC).

FIG. 14 illustrates inductor current and voltage waveforms when theconverter of FIG. 1 and FIG. 12 is operating with reduced outputvoltage. Link inductor 120 current from the input increases during modes1 and 2 to a maximum level as for the full output voltage case of FIG.13, but since the output voltage is half as high as for the full outputvoltage case, link inductor current decreases only half as quickly whiledischarging to the output phases in modes 3 and 4. This will generallysupply the required output current before the link inductor current hasfallen to zero or even near zero, such that there is a significantamount of energy left in the link inductor at the end of mode 4 in FIG.14. This excess energy is returned to the input in mode 5 and 1. Mode 1in FIG. 14 begins prior to the vertical axis. It can be seen that withzero output voltage, the current during modes 3 and 4 (and 7 and 8) willnot decrease at all, so that all link inductor energy is returned to theinput, allowing for the delivery of output current but with no powertransfer, as is required for current delivered at zero volts.

The Kim converter cannot return this excessive inductor energy back tothe input, as this requires bidirectional switches. Thus the Kimconverter must wait until the inductor energy drops to a sufficientlylow value, with the result that the link reactance frequency drops to avery low value as the output voltage approaches zero. This in turn cancause resonances with input and/or output filters. With zero voltageoutput, the Kim converter cannot function at all.

Note that the modes cited in Kim et al. differ somewhat from the modescited here. This is due to two reasons. The first is that, for brevity,the “capacitor ramping”, or “partial resonant” periods in this inventionare not all numbered, as there are 8 of those periods. As indicated inFIGS. 12 b and 12 g, voltage ramping periods preferably occur betweeneach successive pair of conduction triodes. The second reason is thatKim et al. operate their converter such that it draws current from oneinput phase pair per power cycle, and likewise delivers current to onephase pair per power cycle. This results in only two conduction modesper link reactance cycle, since their converter only has one power cycleper link reactance cycle. By contrast, FIG. 12 shows current being drawnand delivered to both pairs of input and output phases, resulting in 4modes for each direction of link inductor current during a power cycle,for a total of 8 conduction modes since there are two power cycles perlink reactance cycle in the preferred embodiment. This distinction isnot dependent on the topology, as either three phase converter may beoperated in either 2 modes or 4 conduction modes per power cycle, butthe preferred method of operation is with 4 conduction modes per powercycle, as that minimizes input and output harmonics. For single phase ACor DC, it is preferred to have only two conduction modes per powercycle, or four modes per link reactance cycle, as there is only oneinput and output pair in that case. For mixed situations, as in theembodiment of FIG. 24 which converts between DC or single phase AC andthree phase AC, there may be 1 conduction mode for the DC interface, and2 for the three phase AC, for 3 conduction modes per power cycle, or 6modes per link reactance cycle. In any case, however, the two conductionmodes per power half-cycle for three phase operation together give asimilar power transfer effect as the singe conduction mode for singlephase AC or DC.

Control algorithms may use this ability of recycling inductor energy toadvantage in order to control current transfers, as is required by manyconverter control algorithms for vector or volts/Hz control. One suchpossible algorithm is explained in FIGS. 16 through 20. FIGS. 16, 17,and 20 show possible current profiles for the link inductor during apower cycle of positive current. This is for the case of only twoconduction modes per power cycle, as this invention uses for singlephase AC or DC. The power cycle for negative inductor current is themirror image of the cycles shown, as there are two power cycles perinductor cycle. Timing intervals T1, T2, T3, Tr1, and Tr2 are shown. T1is the time for the first conduction mode, when current is increasingfrom the input. T2 is the second conduction mode, in which the inductoris connected to the output, either decreasing in current for powertransfer to the output (positive power) as in FIGS. 16 and 17, orincreasing in current for power transfer from the output negative power)as in FIG. 20. T3 is the actually the first part of conduction mode 1 inwhich excess link inductor energy is either returned to the input duringpositive power or delivered from output to input during negative power.Tr1 and Tr2 are the “partial resonant”, or “capacitor ramping” timesduring which all switches are off and the voltage on the link reactanceis ramping. For three phase operation, intervals T1 and T2 aresub-divided, with T1 consisting of two conduction modes for the twoinput phase pairs from which current is drawn, and likewise for T2 fordeliver of current to the output phases. The relative times and inductorcurrent levels determine the charge and therefore the relative currentlevels among the phases. For three phase operation with zero ornear-zero power factor, T2 may subdivided into negative and positiveenergy transfer periods. Note that similar durations are used forramping the converter in BOTH directions. However, the ramping durationscan be different between input and output phases, as load draw variesdue to extrinsic circumstances. The charge time from the input can beheld constant, with the discharge time to the output varied to varyaverage output current (see FIG. 19). Excess link inductor energy(current) is returned to the input in T3. But all charge times andtransitions on the link reactance are perfectly symmetric about the zeropoints of voltage and current (see FIG. 13).

For the single phase AC and DC operation of FIGS. 16 through 20, theaverage output current is given by the equation at the bottom of FIGS.16, 17, and 20, with the “Charge over T2” given by the integral of thelink inductor current over the time interval of T2. For positive power,the peak link inductor current I1 may be held constant, while T2 isvaried, to control average output current: (Iavg-out). An algorithm tocalculate Iavg-out is shown in FIG. 18. For a given set of circuitparameters and input and output voltages, T2 (first column in FIG. 18)may be varied to control Iavg-out (6^(th) column). Resulting other timeintervals and power levels are also calculated. An input voltage of 650volts and an output voltage of 600 volts is used for FIG. 19. FIG. 19shows the results of the algorithm for other output voltages, with the650 volt input, as a function of T2, in micro-seconds (uS). An average(filtered) output current level of 26 amps is shown for the 650 voltoutput curve with a T2 of 27 uS, for a power output of 16.8 kW. Notethat the link reactance frequency remains constant at 10 kHz for the 650volt output curve, regardless of T2 and Iavg-out. For the other curves,with lower output voltage, frequency drops for lower output voltage, butnever drops below 5 kHz even for zero output volts. Also note thatIavg-out for 0 volts goes to 55 amps for T2 of 50 uS, which is more thandouble Iavg-out at maximum power, even though maximum inductor currentremains constant at 110 amps. For lower converter losses when loweroutput currents are commanded, the controller 1500 may be programmed toreduce T1, thereby reducing the peak inductor current.

FIG. 19 also shows sonic specific drive parameters for the example 460VAC, 20 hp drive. The link inductor is 140 μH, and may be constructed asan air core copper wound inductor, with thin, flat, ribbon wire so as tohave a low ratio of AC to DC resistance from the skin effect, and woundlike a roll of tape. This configuration optimizes the inductance toresistance ratio of the inductor, and results in relatively highparasitic capacitance. Such a design cannot be used by hard switchedconverters, as this high parasitic capacitance causes high losses, butwith this invention the high parasitic capacitance is a benefit. Theramp, or parallel, link capacitance is comprised of two parallel AVX(FSV26B0104K-) 0.1 μF film capacitors capable of handling the RMScurrent load of about 25 amps. Peak inductor current is 110 amps.Commercially available reverse-blocking IGBT switches. IXYS part 40N12055 A, 1200 V, arranged in anti-parallel pairs as shown in FIG. 2, 1200,may be used. In a standard hard switched application, such as a currentsource drive, this switch has relatively high turn-on and reverserecovery losses caused by the slow reverse recovery time of the device,but when used in this invention, both turn-on and reverse recoverylosses are negligible even at a per device maximum switching frequencyof 10 kHz and 110 amps peak current. High RMS current capacitors fromAVX (FFV34I0406K), totaling 80 μF line-to-neutral, may be used for theinput and output capacitors. The Altera Cyclone III FPGA may be used forthe controller, implementing the algorithms described above to controlcurrent flow, and using either vector or Volts/Hz to control a 20 hpmotor. Isolated power supplies, gate drivers, and digital isolatorsallow the FPGA to control the on-off states of the IGBTs. Voltage andcurrent sensing circuits, with analog-digital interfaces to the FPGA,allow for precise switch timing to control current flow.

As may be surmised by those skilled in the art, the current resultingfrom the above described operation of the converter is in manyapplications, controlled by controller 1500 to result in a sinusoidalvarying current from the input, normally in phase with the input voltageso as to produce a unity power factor on the input, and sinusoidallyvarying voltage and current on the motor, so as to operate the motor atthe highest possible efficiency and/or performance,

In those cases where the motor is acting as a generator, as may occurwhen the frequency applied to the motor via the converter is rapidlydecreased, the above described operating cycle is reversed, with currentbeing drawn from the motor phases and injected into the input phases.

In general, the input and output frequencies are substantially less thanthe frequency at which the link reactance is operated. For 60 Hz input,a typical operating frequency of the link reactance may be 10 kHz forlow voltage (230 to 690 VAC) drives and converters, and 1.5 kHz formedium voltage (2300 on up) drives and converters, with current pulsefrequencies twice those frequencies, or higher if multiple, synchronizedpower in are used, as shown in FIG. 28. Input and Output frequencies mayvary from zero (DC) to over well over 60 Hz, and may even be up to 20kHz in audio amplifier applications.

The motor drive of FIG. 1 has the following characteristics—

-   -   Low harmonic, unity power factor current draw from the utility,        regardless of output voltage. Current is drawn from each phase        in high frequency pulses, similar to a current source converter,        with input capacitors and optionally, line inductors, converting        the pulsed current flow to sinusoidal current flow.        -   Ability to step up or step down voltage from input to            output, allows full output voltage even in the presence of            input voltage sags, as commonly occurs in industrial power            systems.        -   Sinusoidal output voltage with small voltage ripple allows            standard induction motors, as well as low reactance            synchronous motors, to be used. Output capacitors filter the            pulsed current. Ripple frequency is always high so as to            avoid any resonance problems with input and/or output            filters or reactances.        -   Ability to supply 200% or higher of nominal output current            at low output voltages, indefinitely, as may be advantageous            for starting large inertial loads. With near zero output            voltages, the converter is operated at about half of maximum            frequency, with the inductor first fully charged by the            input, then discharging at that full level into the output            for twice the full voltage discharge period, then            discharging to zero current back into the input, repealing            that cycle but with reverse current. Peak currents remain            the same, but output current is doubled.        -   Input-Output isolation, resulting in zero common mode            voltages on the output. Since there is never a moment when            the input and output lines are connected together, as            happens continuously in voltage and current source drives,            as well as matrix converters, the average output voltage            remains at ground potential. This eliminates the need for            isolation transformers.        -   Slow reverse recovery devices are usable. Rate of change of            current during commutation is relatively slow, and applied            reverse voltage after reverse recovery is also low, so the            switches used may have rectifier diode like recovery            characteristics. Reverse blocking IGBTs and GTOs are            inherently slow to reverse recover, and so this invention is            well suited for these devices.        -   Slower forward turn-off devices are usable. Turn-off dv/dt            is relatively low due to the capacitance in parallel with            the inductor.        -   Compact, lightweight, and efficient. Voltage source drives            with input/output quality similar to this invention require            multiple heavy and bulky power inductors, one on each of the            input and output lines. Current source drives require a very            large and heavy DC inductor in order to generate Inn output            voltage. This invention only needs a single small, compact            AC inductor and the relatively small and lightweight input            and output filter capacitors and input line reactor. Total            weight for a suitably filtered, commercially available            voltage source drive for 40 hp is over 300 pounds, while the            drive of this invention will weigh less than 30 lbs for 40            hp. Lack of large input/output filter inductors            significantly improves the efficiency of this invention over            conventional drives. No transformers are needed since input            current harmonics are low and there is no common node output            voltage.        -   Moderate parts count. Using bi-directional switches, only 12            power switches are needed for this invention. Using            commercially available unidirectional switches with reverse            blocking (reverse blocking IGBT or GTO) requires 24            switches. A 12 pulse input voltage source drive requires 24            switches (18 diodes and 6 active switches).        -   High bandwidth. Since the current amplitude is determined            twice each cycle of the inductor, the current control            bandwidth of this invention is inherently very high, making            the invention suitable for high bandwidth servo applications            and even high power audio amplifiers.

Another embodiment of this invention is shown in FIG. 21, which shows asingle phase AC or DC to single phase AC or DC converter. Either or bothinput and output may be AC or DC, with no restrictions on the relativevoltages. If a portal is DC and may only have power flow either into orout of said portal, the switches applied to said portal ma beuni-directional. An example of this is shown with the photovoltaic arrayof FIG. 23, which can only source power.

FIG. 22 shows an embodiment of the invention as a Flyback Converter. Thecircuit of FIG. 21 has been modified, in that the link inductor isreplaced with a transformer 2200 that has a magnetizing inductance thatfunctions as the link inductor. Any embodiment of this invention may usesuch a transformer, which may be useful to provide full electricalisolation between portals, and/or to provide voltage and currenttranslation between portals, as is advantageous, for example, when afirst portal is a low voltage DC battery bank, and a second portal is120 volts AC, or when the converter is used as an active transformer.

In the embodiments of this invention shown in FIGS. 23 and 24, thenumber of portals attached to the link reactance is more than two,simply by using more switches to connect in additional portals to theinductor. As applied in the solar power system of FIG. 23, this allows asingle converter to direct power flow as needed between the portals,regardless of their polarity or magnitude. Thus, the solar photovoltaicarray may be at full power, 400 volts output, and delivering 50% of itspower to the battery bank at 320 volts, and the 50% to the house AC at230 VAC. Prior art requires at least two converters to handle thissituation, such as a DC-DC converter to transfer power from the solar PVarray to the batteries, and a separate DC-AC converter (inverter) totransfer power from the battery bank to the house, with consequentialhigher cost and electrical losses. The switches shown attached to thephotovoltaic power source need be only one-way since the source is DCand power can only flow out of the source, not in and out as with thebattery.

In the power converter of FIG. 24, as could be used for a hybridelectric vehicle, a first portal is the vehicle's battery bank, a secondportal is a variable voltage, variable speed generator run by thevehicle's engine, and a third portal is a motor for driving the wheelsof the vehicle. A fourth portal, not shown, could be external singlephase 230 VAC to charge the battery. Using this single converter, powermay be exchanged in any direction among the various portals. Forexample, the motor/generator may be at full output power, with 50% ofits power going to the battery, and 50% going to the wheel motor. Thenthe driver may depress the accelerator, at which time all of thegenerator power may be instantly applied to the wheel motor. Conversely,if the vehicle is braking, the full wheel motor power may be injectedinto the battery bank, with all of these modes using a single converter.

FIGS. 25 and 26 show half-bridge converter embodiments of this inventionfor single phase/DC and three phase AC applications, respectively. Thehalf-bridge embodiment requires only 50% as many switches, but resultsin 50% of the power transfer capability, and gives a ripple current inthe input and output filters which is about double that of the fullbridge implementation for a given power level.

FIG. 27 shows a sample embodiment as a single phase to three phasesynchronous motor drive, as may be used for driving a householdair-conditioner compressor at variable speed, with unity power factorand low harmonics input. Delivered power is pulsating at twice the inputpower frequency.

FIG. 28 shows a sample embodiment with dual, parallel power modules,with each module constructed as per the converter of FIG. 1, excludingthe I/O filtering. This arrangement may be advantageously used wheneverthe converter drive requirements exceed that obtainable from a singepower module and/or when redundancy is desired for reliability reasonsand/or to reduce I/O filter size, so as to reduce costs, losses, and toincrease available bandwidth. The power modules are best operated in amanner similar to multi-phase DC power supplies such that the linkreactance frequencies are identical and the current pulses drawn andsupplied to the input/output filters from each module are uniformlyspaced in time. This provides for a more uniform current draw andsupply, which may greatly reduce the per unit filtering requirement forthe converter. For example, going from one to two power modules,operated with a phase difference of 90 degrees referenced to each of themodules inductor/capacitor, produces a similar RMS current in the I/Ofilter capacitors, while doubling the ripple frequency on thosecapacitors. This allows the same I/O filter capacitors to be used, butfor twice the total power, so the per unit I/O filter capacitance isreduced by a factor of 2. More importantly, since the ripple voltage isreduced by a factor of 2, and the frequency doubled, the input linereactance requirement is reduced by 4, allowing the total line reactormass to drop by 2, thereby reducing per unit line reactance requirementby a factor of 4.

FIG. 29 shows an embodiment as a three phase Power Line Conditioner, inwhich role it may act as an Active Filter and/or supply or absorbreactive power to control the power factor on the utility lines. If abattery, with series inductor to smooth current flow, is placed inparallel with the output capacitor 2901, the converter may then operateas an Uninterruptible Power Supply (UPS).

According to various disclosed embodiments, there is provided: ABuck-Boost Converter, comprising: an energy-transfer reactance; a firstbridge switch array comprising at least two bidirectional switchingdevices which are jointly connected to operatively connect at least oneterminal of said reactance to a power input, with reversible polarity ofconnection; a second bridge switch array comprising at least twobidirectional switching devices which are jointly connected tooperatively connect at least one terminal of said reactance to a poweroutput, with reversible polarity of connection; wherein said firstswitch array drives said reactance with a nonsinusoidal voltagewaveform.

According to various disclosed embodiments, there is provided: ABuck-Boost Converter, comprising: an energy-transfer reactance; firstand second power portals, each with two or more ports by whichelectrical power is input from or output to said portals; first andsecond half-bridge switch arrays interposed between said reactance acida respective one of said portals, and each comprising one bidirectionalswitching device for each said port of each said power portal; whereinsaid switch arrays are each operatively connected to respective ones ofsaid portals.

According to various disclosed embodiments, there is provided: AFull-Bridge Buck-Boost Converter, comprising: first and second fullbridge switch arrays, each comprising at least four bidirectionalswitching devices; a substantially parallel inductor-capacitorcombination symmetrically connected to be driven separately by eithersaid switch array; one of said switch arrays being operatively connectedto a power input, and the other thereof being operatively connected tosupply a power output.

According to various disclosed embodiments, there is provided: ABuck-Boost Converter, comprising: first and second switch arrays, eachcomprising at least two bidirectional switching devices; a substantiallyparallel inductor-capacitor combination connected to each said switcharray; wherein a first one of said switch arrays is operativelyconnected to a power input, and is operated to drive power into saidinductor-capacitor combination with a non-sinusoidal waveform; andwherein a second one of said switch arrays is operated to extract powerfrom said inductor-capacitor combination to an output.

According to various disclosed embodiments, there is provided: ABuck-Boost Converter, comprising: first and second switch arrays, eachcomprising at least two bidirectional switching devices; anenergy-transfer reactance connected to each said switch array; wherein afirst one of said switch arrays is connected through respectivecapacitive reactances to a polyphase power input, and operated to drivepower into said reactance from multiple different legs of said powerinput in succession with a non-sinusoidal waveform; and wherein a secondone of said switch arrays is operated to extract power from saidreactance to an output.

According to various disclosed embodiments, there is provided: A powerconverter, comprising: an energy-transfer reactance comprising at leastone inductor; an input switch array configured to drive AC currentthrough said reactance; and an output network connected to extractenergy from said reactance; wherein said input switch array performs atleast two drive operations, in the same direction but from differentsources, during a single half-cycle of said reactance.

According to various disclosed embodiments, there is provided: A powerconverter, comprising: an energy-transfer reactance comprising at leastone inductor, and operating at a primary AC magnetic field frequencywhich is less than half of the reactance's resonant frequency; an inputswitch array configured to drive AC current through said reactance; andan output network switch array connected to extract energy from saidreactance; wherein said input switch array performs at least two driveoperations, in the same direction but from different sources, during asingle half-cycle of said reactance.

According to various disclosed embodiments, there is provided: A powerconverter, comprising: an energy-transfer reactance comprising at leastone inductorr, and operating at a primary AC magnetic field frequencywhich is less than half of the reactance's resonant frequency; an inputswitch array configured to drive current through said reactance; and anoutput switch array to extract energy from said reactance; wherein saidinput switch array performs at least two different drive operations atdifferent times during a single cycle of said reactance, and whereinsaid output switch array performs at least two different driveoperations at different times during a single cycle of said reactance.

According to various disclosed embodiments, there is provided: ABuck-Boost Converter, comprising: an energy-transfer reactancecomprising at least one inductor; an input switch array configured todrive AC current, with no average DC current, through said reactance;and an output network connected to extract energy from said reactance.

According to various disclosed embodiments, there is provided: ABuck-Boost Converter, comprising: an energy-transfer reactancecomprising at least one inductor; a plurality of input switch arrays,each said array configured to drive AC current, with no average DCcurrent, through said reactance; and a plurality of output switcharrays, each connected to extract energy from said reactance; saidarrays having no more than two switches driving or extracting energyfrom said reactance at any given time; wherein said input switch arraysindividually drive said reactance with a nonsinusoidal voltage waveform.

According to various disclosed embodiments, there is provided: A powerconversion circuit, comprising an input stage which repeatedly, atvarious times, drives current into the parallel combination of aninductor and a capacitor, and immediately thereafter temporarilydisconnects said parallel combination from external connections, tothereby transfer some energy from said inductor to said capacitor;wherein said action of driving current is performed in opposite sensesand various times, and wherein said disconnecting operation is performedsubstantially identically for both directions of said step of drivingcurrent; and an output stage which extracts energy from said parallelcombination, to thereby perform power conversion.

According to various disclosed embodiments, there is provided: A powerconversion circuit, comprising: an input stage which repeatedly drivescurrent into the parallel combination of an inductor and a capacitor,and immediately thereafter temporarily disconnects said parallelcombination from external connections, to thereby transfer some energyfrom said inductor to said capacitor; wherein said input stage drivescurrent in different senses at different times; and an output stagewhich repeatedly couples power out of said parallel combination, andimmediately thereafter temporarily disconnects said parallel combinationfrom external connections, to thereby transfer some energy from saidinductor to said capacitor; wherein said output stage couples power outof said combination during two opposite directions of current therein;wherein said input and output stages both disconnect said parallelcombination substantially identically for both directions of current insaid combination.

According to various disclosed embodiments, there is provided: A SoftSwitched Universal Full-Bridge Buck-Boost Converter, comprising: aninductor with a first and second port; a capacitor attached in parallelwith said inductor; connections to a plurality of voltage sources orsinks (portals) of electric power each with a plurality of ports; afirst set of electronic bi-directional switches that comprise saidconnections between said first port of the inductor and each said portof each said portal, with one said switch between the first port of theinductor and each port of each portal; a second set of electronicbi-directional switches that comprise said connections between saidsecond port of the inductor and each port of each portal, with oneswitch between the second port of the inductor and each port of eachportal; capacitive filtering means connected between each said portwithin each said portal; control means to coordinate said switches toconnect said inductor to port pairs on each portal, with no more thantwo switches enabled at any given time; said control means furthercoordinating said switches to first store electrical energy in theinductor by enabling two switches on a given input portal to connect theinductor to said input portal, then disabling the switches after theproper amount of energy has been stored in the inductor; and saidcontrol means may enable further pairs of switches on the same or otherinput portals so as to further energize the inductor, and disable saidswitches after the appropriate inductor energizing is complete; saidcontrol means further enables another pair of switches on another,output, portal to transfer sonic or all of the inductor energy into saidoutput portal, and then disables said switches after the desired amountof charge has been transferred to said portal; said control means mayenable further pairs of switches on the same or other output portals soas to further send charge into said output portals, and disable saidswitches after the desired amount of charge has been transferred to saidportal; and if the inductor has excess energy after discharging into thelast output portal, said control means then enables an appropriateswitch pair to direct said excess energy back into the input portal;wherein said control means may modify the above sequence so as toachieve any required energy transfer among the ports and portals; saidinductor magnetically storing electrical energy in the form of electriccurrent, using said switches; energy transfer from one or more inputportals to said inductor occurring, via current flow through two or moresaid ports of one or more said portals, with only one pair of ports; andcyclically repeating said energy and charge transfers.

According to various disclosed embodiments, there is provided: ASoft-switched Half-Bridge Buck-Boost Converter, comprising: first andsecond power portals, each with two or more ports by which electricalpower is input from or output to said portals, first and secondhalf-bridge switch arras, each comprising one bidirectional switchingdevice for each said port of each said power portal, an energy-transferlink reactance with one port connected to both said switch arrays, andwith the other port connected to an actual or virtual ground, such thatsaid actual or virtual ground maintains at a relatively constantvoltage, each of said switch arrays being connected to a power portalwith said portal possessing capacitive reactance between the legs ofsaid portals configured so as to approximate a voltage source, withpower transfer occurring between said portals via said energy-transferreactance, said link energy-transfer reactance consisting of an linkinductor and capacitance in parallel, said power transfer beingaccomplished in a first power cycle as one or more pairs of input portallegs are singularly or sequentially connected to said energy-transferreactance to store energy via increased current flow and inductance intosaid link inductor, followed by one or more pairs of output portal legssingularly or sequentially connected to said energy-transfer reactanceto remove energy via decreased current flow and inductance from saidlink inductor, with any excess energy in said link inductor subsequentlyreturned back to one or more said input portal leg pairs, followed by areversal of current within said link inductor and a repeat of theheretofore described energy transfer, to constitute a second powercycle, from input to output portal leg pairs, but with opposite butequal current flow in said link inductor and utilizing switches of saidswitch arrays which are complimentary to said switches used for saidfirst cycle of said power transfer; said first and second power cyclescomprising a single voltage cycle of the energy-transfer link reactance;said capacitance, in conjunction with said current reversal, producingsoft-switching of said switches with low-voltage turn-off, zero voltageturn-on, and low reverse recovery losses; said bidirectional switchingdevices being capable of blocking voltage in either direction andconducting current in either direction; wherein said power transfercycles are continuously repeated by said control means to produce saidpower transfer on a continuing basis; and wherein control meanscoordinate said switching actions to produce current and power transfervia said power cycles as required to produce desired output voltage andcurrent, as may be used to drive single or polyphase motors at variablespeed and voltage, or to drive any other electrical DC, single phase AC,polyphase AC, and/or multiple DC loads; said capacitance, in conjunctionwith said current reversal, producing soft-off-switching of saidswitches with low-voltage turn-off, as current is shunted from eachturning-off switch into said substantially parallel capacitance, saidswitches having soft turn-on as diodes as the link reactance voltagecauses control means enabled switches to transition from reverse toforward bias, said switches having soft reverse blocking turn-off as thelink inductor current linearly decreases to zero after discharging intoan output port.

According to various disclosed embodiments, there is provided: ASoft-switched Full-Bridge Buck-Boost Converter, comprising: first andsecond power portals, each with two or more ports by which electricalpower is input from or output to said portals, first and secondfill-bridge switch arrays, each comprising two bidirectional switchingdevices for each said port of each said power portal, a energy-transferlink reactance symmetrically connected to both said switch arrays, eachof said switch arrays being connected to a power portal with said portalpossessing capacitive reactance between the legs of said portalsconfigured so as to approximate a voltage source, with power transferbetween said portals via said energy-transfer reactance, Said linkenergy-transfer reactance consisting of an link inductor and capacitancein parallel, said power transfer being accomplished in a first powercycle as one or more pairs of input portal legs are singularly orsequentially connected to said energy-transfer reactance to store energyvia increased current flow and inductance into said link inductor,followed by one or more pairs of output portal legs singularly orsequentially connected to said energy-transfer reactance to removeenergy via decreased current flow and inductance from said linkinductor, with any excess energy in said link inductor subsequentlyreturned back to one or more said input portal leg pairs, followed by areversal of current within said link inductor and a repeat of theheretofore described energy transfer, to constitute a second powercycle, from input to output portal leg pairs, but with opposite butequal current flow in said link inductor and utilizing switches of saidswitch arrays which are complimentary to said switches used for saidfirst cycle of said power transfer; Said first and second power cyclescomprise a single voltage cycle of the energy-transfer link reactance;Said bidirectional switching devices being capable of blocking voltagein either direction and conducting current in either direction; Saidpower transfer cycles being continuously repeated by said control meansto produce said power transfer on a continuing basis; Said control meanscoordinating said switching actions to produce current and powertransfer via said power cycles as required to produce desired outputvoltage and current, as may be used to drive single or polyphase motorsat variable speed and voltage, or to drive any other electrical DC,single phase AC, polyphase AC, and/or multiple DC loads; Saidcapacitance, in conjunction with said current reversal, producingsoft-off-switching of said switches with low-voltage turn-off, ascurrent is shunted from each turning-off switch into said substantiallyparallel capacitance; Said switches having soft turn-on as diodes as thelink reactance voltage causes control means enabled switches totransition from reverse to forward bias; Said switches having softreverse blocking turn-off as the link inductor current linearlydecreases to zero after discharging into an output port.

According to various disclosed embodiments, there is provided: Anelectric, vehicle, comprising at least one motor, at least oneelectrical energy storage device, and a power converter as above.

According to various disclosed embodiments, there is provided: A solarenergy system comprising at least one photovoltaic array, at least oneelectrical energy storage device, and a power converter as above.

According to various disclosed embodiments, there is provided: A motorsystem comprising a polyphase power line connection, a polyphase motor,and a power converter as above connected therebetween as avariable-frequency drive.

According to various disclosed embodiments, there is provided: Amultiple power module soft-switched converter, comprising multipleconverters as above connected in parallel between an input portal and anoutput portal, and commonly controlled to minimize harmonics in thecurrent drawn from and delivered to said input and output portals.

According to various disclosed embodiments, there is provided: Accordingto various disclosed embodiments, there is provided: A composite of nconverters as above, connected at least partially in parallel, andoperating at inductor phase angles separated by 180/n degrees; wherebythe amount of input/output filtering can be reduced.

According to various disclosed embodiments, there is provided: A methodfor operating a Buck-Boost Converter, comprising the actions of: (a)operating a first bridge switch array, comprising bidirectionalswitching devices, to operatively connect at least one terminal of areactance to a power input, with polarity which reverses at differenttimes; (b) operating a second bridge switch array, comprisingbidirectional switching devices, to operatively connect at least oneterminal of said reactance to a power output, with polarity whichreverses at different times; wherein said actions (a) and (b) are neverperformed simultaneously.

According to various disclosed embodiments, there is provided: A methodfor operating a Buck-Boost Converter, comprising the actions of:operating a first bridge switch array, comprising bidirectionalswitching devices, to operatively connect at least one terminal of asubstantially parallel inductor-capacitor combination to a power input,with polarity which reverses at different times; wherein said firstswitch array is operatively connected to a power input, and is operatedto drive power into said inductor-capacitor combination with anon-sinusoidal waveform; and operating a second one of said switcharrays to extract power from said inductor-capacitor combination to anoutput.

According to various disclosed embodiments, there is provided: A methodfor operating a power converter, comprising the actions of: driving anenergy-transfer reactance with a full AC waveform, at a base frequencywhich is less than half the resonant frequency of said reactance;coupling power into said reactance, on each cycle thereof, with twodifferent drive phases, respectively supplied from two different legs ofa polyphase power source; and coupling power out of said reactance, oneach cycle thereof, with two different connection phases, respectivelydriving two different legs of a polyphase power output.

According to various disclosed embodiments, there is provided: A methodfor power conversion, comprising the actions of: driving, anenergy-transfer reactance with a full AC waveform, at a base frequencywhich is less than half the resonant frequency of said reactance;coupling power into said reactance, on each cycle thereof, with twodifferent drive phases, respectively supplied from two different legs ofa polyphase power source; and extracting power from said reactance to anoutput.

According to various disclosed embodiments, there is provided: ABuck-Boost power conversion method, comprising: operating an inputswitch array configured to drive AC current through an energy-transferreactance, at an average current magnitude which is more than 100 timesas great as the average DC current within said reactance; saidenergy-transfer reactance comprising at least one inductor; andoperating an output network to extract energy from said reactance.

According to various disclosed embodiments, there is provided: A methodfor operating a power conversion circuit, comprising the steps ofrepeatedly, at various times: driving current into the parallelcombination of an inductor and a capacitor, and immediately thereaftertemporarily disconnecting said parallel combination from externalconnections, to thereby transfer some energy from said inductor to saidcapacitor; wherein said action of driving current is performed inopposite senses and various times, and wherein said disconnectingoperation is performed substantially identically for both directions ofsaid step of driving current; and extracting energy from said parallelcombination, to thereby perform power conversion.

According to various disclosed embodiments, there is provided: A methodfor operating a power conversion circuit, comprising the steps ofrepeatedly, at various times: a) driving current into the parallelcombination of an inductor and a capacitor, and immediately thereaftertemporarily disconnecting said parallel combination from externalconnections, to thereby transfer some energy from said inductor to saidcapacitor; b) coupling power out of said parallel combination, andimmediately thereafter temporarily disconnecting said parallelcombination from external connections, to thereby transfer some energyfrom said inductor to said capacitor; wherein said disconnectingoperation, in said step a, is performed substantially identically forboth directions of said step of driving current; and wherein saiddisconnecting operation, in said step b, is performed substantiallyidentically for both directions of said step of driving current.

According to various disclosed embodiments, there is provided: Methodsand systems for transforming electric power between two or more portals.Any or all portals can be DC, single phase AC, or multi-phase AC.Conversion is accomplished by a plurality of bi-directional conductingand blocking semiconductor switches which alternately connect aninductor and parallel capacitor between said portals, such that energyis transferred into the inductor from one or more input portals and/orphases, then the energy is transferred out of the inductor to one ormore output portals and/or phases, with said parallel capacitorfacilitating “soft” turn-off, and with any excess inductor energy beingreturned back to the input. Soft turn-on and reverse recovery is alsofacilitated. Said bi-directional switches allow for two power transfersper inductor/capacitor cycle, thereby maximizing inductor/capacitorutilization as well as providing for optimum converter operation withhigh input/output voltage ratios. Control means coordinate the switchesto accomplish the desired power transfers.

Modifications and Variations

As will be recognized by those skilled in the art, the innovativeconcepts described in the present application can be modified and variedover a tremendous range of applications, and accordingly the scope ofpatented subject matter is not limited by any of the specific exemplaryteachings given. It is intended to embrace all such alternatives,modifications and variations that fall within the spirit and broad scopeof the appended claims.

While the proceeding Figures illustrate exemplary embodiments of aconverter, Buck-Boost converter and methods of operation therefore,other circuits (including variations of the foregoing circuits) andmethods of operation therefore are well within the broad scope of thepresent invention. For a better understanding of power electronicsincluding Buck-Boost converter technologies, see Principles of PowerElectronics, by Kassakian, M. Schlecht, Addison-Wesley PublishingCompany (1991). The aforementioned reference is herein incorporated byreference.

The disclosed converter circuits are advantageously applicable to a widevariety of systems, including for example:

-   -   Electric vehicles, in which electrical interconversion is        required among some or all of a traction motor, a battery, an        energy source (engine or fuel cell), and an external charging        connection. The source impedances and load impedances of all        these elements can be very different from each other, and can        vary widely over time with different load conditions or        hysteretic history. Moreover, the traction motor itself can be        operated, using the disclosed converter technology, as a        variable-frequency AC drive.    -   Photovoltaic systems, as discussed above, are another attractive        application. Here too electrical interconversion is required        among some or all of a photovoltaic array, a battery array, a        utility input, an energy source (engine or fuel cell), unknown        line loads (applicances), and possibly an external power filter        with significant stored energy. In this application reactive        power compensation may also be desired.    -   Variable-frequency motor drive is an attractive and extremely        broad class of applications. Note that online systems according        to the present application can also be used for reactive power        compensation, and/or to implement soft shutdown using a stored        energy source. Online systems according to the present        application can also be easily reconfigured for a very wide        variety of source or power line voltages and frequencies,        possibly with a change of inductor and/or a change of switches.        Motor-generator traction applications can particularly benefit        from less stringent requirements on generator power quality.    -   HVDC transmission is another attractive class of applications.        In this case the reduced requirements for switch ratings are        particularly attractive.    -   Large arc and plasma drive applications are also very        attractive. In such cases the load often has a negative marginal        imipedance, and active current control is very useful. In many        applications, such as arc furnaces, the impedance of the load        may change substantially as a process progresses, and the agile        control capabilities of the disclosed system configurations can        be very advantageous here.    -   In general, the very high bandwidth active control ability of        the disclosed inventions are useful in a wide range of systems.        The disclosed converter architectures are much better, in this        respect, than current-source converters, and even than        voltage-source converters.

Additional general background, which helps to show variations andimplementations, may be found in the following publications, all ofwhich are hereby incorporated by reference:

-   -   U.S. Pat. Nos. 5,903,448, 4,616,300, 6,404,654, 5,977,569, and        7,057,905;    -   Ngo, “Topology and Analysis in PWM Inversion, Rectification, and        Cycloconversion” Dissertation (1984);    -   Kim and Cho, “New Bilateral Zero Voltage Switching AC/AC        Converter Using High Frequency Partial-resonant Link”, IEEE        (1990);    -   K. Rajashekara et al., “Power Electronics”, Chapter 30 of The        Electrical Engineering Handbook (ed. R. Dorf 2000); M.        Kassakian, Principles of Power Electronics, (1991).    -   M. Brown, Practical Switching Power Supply Design (1990);    -   Cheron: Soft Commutation (1992);    -   Facts Worth Knowing about Frequency Converters 2ed. (Danfoss)        (1992);    -   Gottlieb, Irving: Power Supplies, Switching Regulators,        Inverters, and Converters (2.ed. 1994);    -   Hughes: Electric Motors and Drives 2ed. (1993)’    -   Kenjo: Power Electronics for the Microprocessor Age 2ed. (1994);    -   Kislovski et al.: Dynamic Analysis of Switching-Mode DC/DC        Converters (1991);    -   Lenk: Simplified Design of Switching Power Supplies (1995);    -   McLyman, C. W. T.: Designing Magnetic Components for High        Frequency DC-DC Converters (1993);    -   Mohan; Power Electronics: Converters, Applications, and Design        2ed. (1995);    -   Nave, Mark: Power Line Filter Design for Switched-Mode Power        Supplies (1991);    -   Schwarz: Design of Industrial Electric Motor Drives (1991);    -   Shah, Rajesh J.: Simplifying Power Supply Tech (1995);    -   Tihanyi, Laszlo: Electromagnetic Compatibility in Power        Electronics (1995);    -   Wu, Keng. C.: Pulse Width Modulated DC-DC Converters 1997).

None of the description in the present application should be read asimplying that any particular element, step, or function is an essentialelement which must be included in the claim scope: THE SCOPE OF PATENTEDSUBJECT MATTER IS DEFINED ONLY BY THE ALLOWED CLAIMS. Moreover, none ofthese claims are intended to invoke paragraph six of 35 USC section 112unless the exact words “means for” are followed by a participle.

The claims as filed are intended to be as comprehensive as possible, andNO subject matter is intentionally relinquished, dedicated, orabandoned.

1-80. (canceled)
 81. A power converter, comprising: a parallelinductor/capacitor combination, having first and second terminals; afirst external power connection, reversibly connected to said first andsecond terminals through respective phase legs each comprising twobidirectional switches; a second external power connection, reversiblyconnected to said first and second terminals through respective phaselegs each comprising two bidirectional switches; and control circuitrywhich, when said inductor/capacitor combination has been connected toone of said external power connections, completely disconnects saidinductor/capacitor combination from all external power connectionsbefore connecting said inductor/capacitor combination to another of saidexternal power connections.
 82. The converter of claim 81, wherein saidfirst external power connection is reversibly connected to said firstand second terminals through a plurality of phase legs, each said phaseleg including a first bidirectional switch which connects a respectiveline of said external power connection to said first terminal, and asecond bidirectional switch which connects said respective line to saidsecond terminal.
 83. The converter of claim 81, wherein said secondexternal power connection is reversibly connected to said first andsecond terminals through a plurality of phase legs, each said phase legincluding a first bidirectional switch which connects a respective lineof said second external power connection to said first terminal, and asecond bidirectional switch which connects said respective line to saidsecond terminal.
 84. A power converter, comprising: a transformershunted by a capacitor; a first external power connection, reversiblyconnected to a first and a second terminal of said transformer throughrespective phase legs each comprising two bidirectional switches, butnot to a third nor a fourth terminal thereof; a second external powerconnection, reversibly connected to said third and fourth terminalsthrough respective phase legs each comprising two bidirectionalswitches, but not to said first and second terminals; and controlcircuitry which, when said transformer has been connected to one of saidexternal power connections, completely disconnects the combination ofsaid transformer and said capacitor from all external power connectionsbefore connecting said combination to another of said external powerconnections.
 85. The converter of claim 84, wherein said first externalpower connection is reversibly connected to said first and secondterminals through a plurality of phase legs, each said phase legincluding a first bidirectional switch which connects a respective lineof said external power connection to said first terminal, and a secondbidirectional switch which connects said respective line to said secondterminal.
 86. The converter of claim 84, wherein said second externalpower connection is reversibly connected to said first and secondterminals through a plurality of phase legs, each said phase legincluding a first bidirectional switch which connects a respective lineof said second external power connection to said first terminal, and asecond bidirectional switch which connects said respective line to saidsecond terminal.